This invention comprises a low noise amplifier capable of handling large signals without significant distortion. The amplifier is particularly useful as the first electronic stage in systems using piezoelectric signal sources although it is not restricted to such signal sources.

In order to measure weak signals, it is important to have low noise levels. Large unwanted signals are frequently encountered along with the signals desired to be detected. Attempts to eliminate these unwanted signals, prior to amplification, will often result in deterioration in the ultimate signal-to-noise ratio. It is preferable, if at all possible, to amplify the entire signal in a linear fashion, and then filter. Thus, it is desirable for a preamplifier to have a large signal handling capability as well as low-noise characteristics. In addition, such a preamplifier should have a gain which is very stable with respect to changes in supply voltage, time and frequency.

Because of the linearity requirement and the stability objective, a feedback design is employed in the present invention with a high quality junction-type field effect input transistor (FET), this having been found to be the most suitable for high impedance, low-frequency applications.

Although the amplifier according to the invention is DC coupled, it is intended to be usable to an arbitrarily low frequency, rather than simply as a DC amplifier. The DC performance is relatively poor by the usual standards for operational amplifiers, but it is entirely adequate for the intended purposes of this amplifier.

In accordance with this invention there is provided an amplifier comprising an input stage, a second stage and a third stage, said input stage comprising a field effect transistor (FET) connected in a common source configuration and having a gate electrode to which is connected an input terminal, said input terminal being connected to a pair of oppositely poled reverse-biased input protection diodes, said field effect transistor (FET) acting as a voltage-to-current converter and feeding said second stage, said second stage comprising a junction transistor connected in a common emitter configuration and acting as a current-to-voltage converter, said third stage comprising a further junction transistor connected as an emitter follower, overall gain of the amplifier being controlled by a feedback connection from the third stage to the first stage, said second and third stages being powered by a bipolar power supply, said second stage junction transistor having an emitter connected to positive supply voltage, said third stage including a second field effect transistor (FET) connected in series with said further junction transistor to enhance output drive capability, said further junction transistor having a base and a collector with resistors connected in series with them to provide output short circuit protection.

The design and operation of the circuit will be described in greater detail, with reference to the following figures, in which:

FIG. 1 is a schematic diagram of the amplifier according to the invention;

FIG. 2 is a typical curve of the open loop gain versus frequency of the amplifier;

FIG. 3 is a typical frequency response curve (closed loop gain) for the amplifier;

FIG. 4 indicates typical noise level curves for the amplifier;

FIG. 5 indicates typical rejection (of power supply noise) curves;

FIG. 6 is a curve of maximum sinewave voltage output versus power supply voltage.

Referring to FIG. 1, the input FET Q1, protected by diodes D1 and D2, acts as a voltage-to-current converter, current feeding the second stage. The second stage includes a junction transistor Q2 connected as a common-emitter current-to-voltage converter for generating an output across R4. In this embodiment, Q2 is a PNP transistor. The current amplification in the second stage is the major source of gain in the amplifier. The third stage is simply an emitter follower Q3 connected in series with Q4 which acts as a current source. The output of the amplifier is taken from the emitter of NPN transistor Q3. Overall gain of the amplifier is determined by the values of the feedback resistors R2 and R7. R1, together with diodes D3 and D4, provides static bias for the second stage. The capacitor C2 ensures a low input impedance for this stage, helping to divert the dynamic current through Q2 instead of R1. The Miller capacitor C1 restricts the gain-bandwidth product of the amplifier, ensuring stability. Resistors R5 and R6 in series with the base and emitter of Q3, respectively, are for current limiting in case of an output short circuit. The capacitors C3 and C4 are for supply line decoupling. A reasonable thermal stability is obtained despite the unbalanced design, and the design has been found to be very successful.

The input stage is a common source FET amplifier comprising FET Q1 which has a source resistor R2 and a drain load which is equivalent to R1 in parallel with the input impedance of the second stage. R2 has a degenerative effect, reducing the effective transconductance g_m ¹ of the stage to g_m /(1+g_m R2) where g_m is the actual transconductance of the FET at operating bias. For noise considerations, R2 should be kept as small as reasonably possible. In addition to contributing noise directly, the DC drop across R2 biases Q1 so as to reduce conductance. With R4 having a typical value of about 360 ohms, g_m ¹ is about 40% of the zero bias g_m. The drain load will typically be about 4000 ohms and the voltage gain of the input stage will be about 5 in the absence of feedback through R7.

The second stage is a common emitter stage comprising junction transistor Q2 and associated passive elements. The forward biased diodes, D3 and D4 together with the base-emitter drop of Q2, allow the DC voltage across R1 to stand at about 2 volts. The absence of diodes would reduce the value of R1 by about a factor of 3 and hence reduce the gain of the first stage and shunt a considerable portion of the available signal current to the positive supply rail 10. The diodes themselves serve no dynamic role and hence are heavily bypassed by capacitor C2. The input impedance h_ie of Q2 is approximately 0.05β/i where β is the current gain of Q2 and i is the collector current. The DC voltage across R4 must be about 16 volts. Hence i=16/R4 and h_ie =0.5βR4/16. The collector signal current is given by βe₂ R4/h_ie and hence the output signal voltage e₃ is βe₂ R4/h_ie, where e₂ is the signal voltage across R1. The second stage voltage gain is thus approximately 16βR4/0.05βR4≈320. In actual circuits the gain has been found to be slightly higher at about 440. The overall gain for the first two stages will be about 2,200.

At frequencies above about 100 Hz, C1 acts as a miller feedback capacitor, reducing both h_ie and the voltage gain. The high frequency voltage gain of the first two stages is given by g_m ¹ XC1 where XC1 is the reactance of C1. For a value of 33 pf, the typical gain-bandwidth product is about 5 MHz. It is important to the stability of the overall circuit that the gainbandwidth product be less than the f_T of the transistors Q2 and Q3.

Resistor R1 is adjusted in each amplifier so as to bias Q2 for zero voltage at the amplifier output.

The two diodes in the emitter of Q2 could be replaced with a resistor of about 2.9 K-ohms. The use of a resistor causes a degenerative effect at low frequencies reducing the amplifier gain to about 80 at DC. This substitution will also result in an overload recovery time of about 3 seconds instead of 30 msec.

The third stage comprises a junction transistor Q3 connected as an emitter follower. Field effect transistor Q4 in series with Q3 is a current source used to "pull down" the output without loading it. The operating current through Q4 must be sufficient to drive R7 plus the external load to -13 volts if the full output potential of the amplifier is to be realized. Resistors R5 and R6 serve to limit base and collector currents in the event of an output short circuit.

Input protection is provided by diodes D1 and D2, which can be a pair of FD300 diodes. Over-voltage protection may be obtained either by connecting these diodes to ground or to a bias source such as a pair of mercury cells. If the diodes are connected to ground, an additional resistance (greater than 13 megohms) will be placed across R3, lowering the input impedance of the amplifier. This additional resistance represents the dynamic zero-bias resistance of the diodes themselves and as such will be a nonlinear function of input level. In order to reduce the input loading while still protecting the amplifier, the diodes must be back-biased by about 1 to 1.5 volts. Small mercury cells can work well here. A resistor chain may be used for bias provided the diode terminals are heavily bypassed to ground with capacitors.

The diodes alone may not be sufficient protection in some cases, e.g. when the signal source is a piezoelectric hydrophone. Pyroelectric charges in piezoelectric hydrophones may reach kilovolt levels. The connection of a charged hydrophone to a preamplifier may result in surge currents of many amperes. Tests with these amplifiers indicated that surge currents up to about five amperes could be tolerated. On this basis, a resistor can be added in series with the input to limit surge current to five amperes or less. The resistor value should be about one ohm for every five volts of protection required. If 2000 volt transients are possible, a resistor of at least 400 ohms should be used. This series resistor will cause a deterioration in noise performance, particularly at frequencies greater than a few hundred hertz.

The open loop gain of an amplifier in accordance with the invention is shown in FIG. 2. The droop at low frequencies is caused by the increasing impedance of C2 as frequency falls.

The closed loop gain is controlled by R7 and R2, and is given approximately by R2/(R2+R7). The closed loop gain is shown in FIG. 3.

The open circuit and short circuit noise spectra are shown in FIG. 4. The open circuit noise is simply that noise associated with the input impedance. A further resistor in series with the input will necessarily increase the short circuit noise by the noise power associated with that resistor. The protection diodes, D1 and D2, were not found to contribute noticeably to the noise with a 10 megohm input resistor R3.

In the absence of obvious clipping, the harmonic distortion would seem to be unmeasurable below 20 KHz. Where it can be measured, total harmonic distortion appears to obey the formula 1.5×10^-5 Vf2% where V is the rms output voltage and f is the frequency in kilohertz. Below 1 KHz, distortion probably levels off at 1.5×10^-5 V%.

Rejection of power supply noise as a function of frequency is shown in FIG. 5. These figures are referred to the input. One millivolt of supply noise looks like less than 1 volt of input noise below 5 KHz.

Maximum output voltage swing as a function of power supply voltage is shown in FIG. 6. It is assumed that the supplies are balanced with respect to ground. Below ±4.5 volts, the amplifier continues to function but gain is reduced. The amplifier is intended for the power supply range ±12 to ±16 volts.

The preamplifier of this invention is quite easily constructed in a modular, plug-in form, by modern manufacturing techniques for integrated circuits. It is to be noted that many modifications to the basic circuit of the amplifier presented herein are possible, and are contemplated by this invention. For instance, different transistor types may be used, the gain may be changed, and impedance levels and bandwidths may be changed.

The following components have been used in an actual amplifier according to the invention and the curves of FIGS. 2 and 6 relate to an amplifier using these components. However, it will be obvious to those skilled in the art that other components could be used.

Q1--2N5556

Q2--2N4250

Q3--2N916

Q4--2N819

D1,D2--FD300

D3,D4--IN914

C1--33 pf

C2--330 μf

C3--1 μf

C4--1 μf

R1--(varies--see text)

R2--360Ω

R3--10 MΩ

R4--47 kΩ

R5--100Ω

R6--100Ω

R7--3.3 kΩ

It will be appreciated that a complementary design may also be used. That is, Q1 may be replaced by a P-channel FET, Q2 by an NPN junction transistor and Q3 by a PNP junction transistor, with diodes D3 and D4 being reversed and the positive and negative voltage supplies being interchanged. Q4 may be either P-channel or N-channel provided it is appropriately biased. It will also be obvious that diodes D3 and D4 may be replaced by any suitable number of forward biased diodes or by one or more back-biased zener or avalanche diodes without altering the basic design.

The amplifier is preferably constructed in a modular plug-in form. The circuit components may be assembled on a printed circuit board and then encapsulated. The resulting unit is relatively small and preferably has pin designations indelibly printed on it. The colour of the encapsulant may be used as a code to indicate the value of input impedence R3.

If desired a pin may be provided which connects to the junction of R2 and R7 so that an external trim resistor may be connected between said pin and the output. The external trim resistor enables adjustment of the overall gain of the amplifier to some lower value.