BACKGROUND AND PRIOR ART

&null;0002&null; A PFC converter is necessary for many electronic equipments to meet harmonic regulations and standards. For low power applications, single stage PFC converter is a better choice considering cost and performance.

&null;0003&null; Typical single-stage PFC topologies with tight output voltage regulation were proposed in the paper by Redl, R.; Balogh, L.; Sokal, N. O., &null;A New Family of Single-Stage Isolated Power-Factor Correctors with Fast Regulation of the Output Voltage&null;, PESC '94, P1137-1144, vol. 2, June 1994. In those topologies, a PFC cell is integrated with a Direct Current/Direct Current (DC/DC) conversion cell, and both cells share active switches and controller, but suffer from high intermediate bus voltage and high current stresses.

&null;0004&null; Some methods to reduce the intermediate bus voltage were discussed in the paper by Qiao, Chongming' Smedley, K. M., &null;A Topology Survey of Single-Stage Power Factor Corrector with a Boost Type Input-Current-Shaper&null; APEC 2000, P460-467, Vol. 1, March 2000. Unfortunately, those methods brings high distortion to the line current waveform, resulting in reduced power factor.

&null;0005&null; One approach to limit intermediate bus voltage was proposed in the publication by Luo, et al., &null;Flyboost Power Factor Correction Cell and Its Applications in Single-Stage AC-DC Converters&null;, PSEC, 2002. By adding a secondary winding to the boost inductor, there was provided two discharging paths for boost inductor: to the intermediate storage capacitor or directly to the load. It means that some input power is directly transferred to the load without being processed by DC/DC conversion cell, referred as parallel power transfer in both the publications by: Jiang, Y.; Lee, F. C.; Hua, G.; Tang, W., &null;A Novel Single-Phase Power Factor Correction Scheme&null;, APEC '93, P287-292, March 1993; and Garcia, O.; Cobos, J. A.; Prieto, R.; Alou, P.; Uceda, J.; &null;Power Factor Correction:A Survey&null;, PESC '01, P8-13 vol. 1, June 2001.

&null;0006&null; These approaches limit the intermediate bus voltage with little influence on input current waveform, and allow the DC/DC conversion cell to operate in a continuous conduction mode (CCM) without high voltage punishment at light load conditions.

&null;0007&null; Various patents have been proposed in this area but fail to overcome all the problems of the prior art.

&null;0008&null; A search was also carried out with the following results:

&null;0009&null; U.S. Pat. No. 5,146,394 to Ishii, et al discloses the use of only one power transformer; U.S. Pat. No. 5,796,595 to Cross has two transformers that are not connected in series and is not for Power Factor Correction applications; U.S. Pat. No. 5,909,361 to Kim discloses the use of only one power transformer; U.S. Pat. No. 5,982,638 to Tang, et al discloses the use of only one power transformer; U.S. Pat. No. 6,005,782 to Jain, et al. discloses the use of only one transformer to deliver the energy to the output and is not for Power Factor Correction applications; U.S. Pat. No. 6,031,747 to Ilic, et al, has no transformer, does not work as a Power Factor Correction cell and is not for Power Factor Correction applications; U.S. Pat. No. 6,115,267 to Herbert wherein the transformer does not work in the Flyback mode and is and without an intermediate DC bus capacitor; U.S. Pat. No. 6,272,027 B1 to Fraidlin, et al has only one transformer and no intermediate DC bus capacitor; U.S. Pat. No. 6,281,666 B1 to Tressler, et al which is not for Power Factor Correction applications and has no intermediate DC bus capacitor; and, U.S. Pat. No. 6,282,109 B1 to Fraidlin, et al has no isolation, no Flyback transformer nor an intermediate DC bus capacitor.

&null;0010&null; Thus, the need exists for a lower cost, lower bus voltage DC/DC conversion cell that can operate in Continuous Conduction Mode (CCM).

SUMMARY OF THE INVENTION

&null;0011&null; It is a primary objective of the present invention to develop a power factor correction topology which limits the intermediate bus voltage to less than 400 volts for universal voltage applications.

&null;0012&null; Another object of this invention is to provide a PFC topology which provides two discharge paths for the choke inductor whereby the intermediate bus voltage has reduced influence on the input current waveform.

&null;0013&null; A further object of this invention is to provide a low cost low power AC/DC converter power supply.

&null;0014&null; A preferred embodiment of the integrates two flyback topologies to enable switching with only a single switch while also requiring only a single capacitor for all universal voltage applications under 400 volts.

&null;0015&null; Further objects and advantages of this invention will be apparent from the following detailed description of presently preferred embodiments which are illustrated schematically in the accompanying drawings.

BRIEF DESCRIPTION OF THE FIGURES

&null;0016&null; FIG. 1(a) is a schematic of the basic topology of the Bi-flyback PFC converter.

&null;0017&null; FIG. 1(b) schematic of the Bi-flyback PFC converter with a low loss snubber circuit.

&null;0018&null; FIG. 1(c) is a schematic of the bi-flyback PFC converter with an active clamp circuit.

&null;0019&null; FIG. 2 shows the operation mode in one line period.

&null;0020&null; FIG. 3(a) shows the operation waveforms resulting from operation of the flyback mode

&null;0021&null; FIG. 3(b) shows the equivalent circuit during the ON period in operation of the flyback mode

&null;0022&null; FIG. 3(c) shows the equivalent circuit during the OFF period in operation of the flyback mode

&null;0023&null; FIG. 4(a) shows the operation waveforms resulting from operation of the boost mode

&null;0024&null; FIG. 4(b) shows the equivalent circuit during the ON period in operation of the boost mode

&null;0025&null; FIG. 4(c) shows the equivalent circuit during the OFF period in operation of the boost mode

&null;0026&null; FIG. 5 shows the power flow over a line period.

&null;0027&null; FIG. 6 illustrates the input current and input voltage at 100 W output and 110V input.

&null;0028&null; FIG. 7 graphs the measured intermediate bus voltage versus output power.

&null;0029&null; FIG. 8 graphs the measured power factor at a 150 W load.

&null;0030&null; FIG. 9 graphs the measured efficiency at a 150 W load.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

&null;0031&null; Before explaining the disclosed embodiments of the present invention in detail, it is to be understood that the invention is not limited in its application to the details of the particular arrangements shown since the invention is capable of further embodiments. Also, the terminology used herein is for the purpose of description and not of limitation

&null;0032&null; According to this invention, the above objects can be achieved by a single stage Bi-flyback Power Factor Correction (PFC) topology. The advantages and features of the present invention will be apparent upon consideration of the following description.

&null;0033&null; The Bi-flyback topology consists of two flyback circuits, as shown in FIG. 1(a). The first PFC flyback circuit is composed of transformer T1 120, rectifier input bridge 140, diode D1 160, output filter capacitor Co 180 and Power Metal Oxide Silicon Field Effect Transistor (MOSFET) Switch (S) 200. The second Direct Current/Direct Current (DC/DC) flyback circuit includes transformer T2 220, intermediate bus capacitor CS 240, diode D2 260, output filter capacitor Co 180 and the same Power MOSFET Switch (S) 200. The PFC cell operates under the Discontinuous Conduction Mode (DCM) and the DC/DC conversion cell operates under a Conduction Mode of the class of the Continuous Conduction Mode (CCM) and DCM during the entire line period.

&null;0034&null; The above referenced second DC/DC flyback circuit operates similar to a DC/DC flyback circuit. For the first PFC flyback circuit, there are two discharging paths for transformer T1 120, depending on instantaneous input voltage and intermediate bus voltage. When input voltage is low, T1 120 will discharge its magnetizing energy to the load 280 via diode D1 160 during S 200 OFF period, like the typical flyback transformer. When input voltage is high, T1 120 operates like a boost inductor and discharges its magnetizing energy to intermediate bus capacitor 240 through T2's 220 primary winding.

&null;0035&null; When T1 120 operates like flyback transformer, it is referred to as the flyback mode, and as the boost mode when T1 120 works as boost inductor. So there are two operational modes for this topology. The operational modes over one line cycle are shown in FIG. 2.

&null;0036&null; In FIG. 1(b), one low loss snubber circuit, comprised of diode 222 and diode 224, capacitor 228, and inductor 226, is added to basic Bi-flyback topology to absorb the voltage spike across the main switch S when it is turned off. In FIG. 1(c), one active clamp ciruit, comprised of capacitor 232 and MOSFET switch s 234, is used to carry out snubber function and zero voltage switching for the main switch.

&null;0037&null; A. Flyback Mode

&null;0038&null; When rectified line voltage &null;Vin(t)&null; is less than VCS&null;n2Vo-n1Vo(n1: the turn ratio of T1 120, n2: the turn ratio of T2 220), Transformer T1 120 works like a flyback transformer, and the topology operates like two independent flyback converters. All input power during this mode is directly transferred to the load 280 through T1 120. Meanwhile, the DC/DC flyback cell will deliver some power from intermediate capacitor 240 to load 280, in order to keep tight output voltage regulation. The equivalent circuits and operational waveforms are shown in FIG. 3.

&null;0039&null; Interval 1 (t0&null;t1): Switch S 200 is turned on at to. The rectified line voltage &null;Vin(t)&null; is applied to primary winding of T1 120. The current in T1 120, i1 in FIG. 3, increases linearly. The intermediate bus voltage VCS is applied to primary winding of transformer T2 220, causing its current i2 to linearly increase also. Since PFC cell operates in DCM, i1 starts increasing from zero.

&null;0040&null; Interval 2 (t1&null;t2): S 200 is turned off at t1. T1 120 discharges through its secondary winding and deliveries stored magnetizing energy to the load 280. The current (iD1) in T1 120 secondary winding decreases linearly. T2 220 also discharges its magnetizing energy to the load 280 via its secondary winding. The voltage across transformer T1 120 primary winding is n1Vo, while the voltage across T2 220 primary winding is n2Vo. So the voltage across switch S 200, VDS, is equal to VCS&null;n2Vo, which is higher than &null;Vin(t)&null;&null;n1Vo. The input rectifier bridge 140 is blocked.

&null;0041&null; Interval 3 (t2&null;t3): At t2, all magnetizing energy in T1 120 is transferred to the load 280. Current iD1 reaches zero and diode D1 160 keeps it at zero. The current in T2 220 secondary winding, iD2, continues to decrease until the switch 200 is turned on at t3. Then a new switching cycle begins. Switching period TS is equal to t3-t0.

&null;0042&null; B. Boost Mode

&null;0043&null; When the line voltage goes higher, &null;Vin(t)&null;>VCS&null;n2Vo&null;n1Vo, the voltage in transformer T1 120 primary winding during S 200 OFF period is VDS&null;&null;Vin(t)&null;, i.e. VCS&null;n2Vo&null;Vin(t)&null;, which will be less than n1Vo. It means that diode D1 160 in T1 120 secondary winding discharge path will not conduct. T1 120 works like a boost inductor and discharges its magnetizing energy only through its primary winding. At this mode will operate as BIFRED topology. T1 120 transfers some input power to intermediate storage capacitor CS 240 and some input power to the load 280 through T2 220. Meanwhile, T2 220 will also transfer some power from intermediate capacitor 180 to the load 280 in order to keep tight output voltage regulation. The equivalent circuits and operational waveforms are shown in FIG. 4.

&null;0044&null; Interval 1 (to&null;t1): S 200 is turned on at to, resulting in rectified line voltage &null;Vin(t)&null; applied to T1 120. The current in T1 120 primary winding, i1 in FIG. 4, increases linearly. And the voltage across the intermediate bus capacitor 180, VCS, is applied to T2 220 primary winding, which causes the current i2 in FIG. 4 to linearly increase also.

&null;0045&null; Interval 2 (t1&null;t2): Switch S 200 is turned off at t1. T1 120 current i1 will decrease linearly and discharge its magnetizing energy through DC/DC transformer T2 220 primary winding and intermediate capacitor 240. So T2 220 primary winding current i2 is equal to T1 120 current i1, which will be reflected to T2 220 secondary winding. The current (iD2) in T2 220 secondary winding consists of magnetizing current of T2 220 and reflected current of T1 120 current i1. And T2 220 magnetizing current will decrease linearly since output voltage is applied to T2 220 secondary winding. The voltage across T2 220 primary winding is n2Vo. So the voltage across S, VDS, is equal to VCS&null;n2Vo. The voltage across transformer T1 120 primary winding is VCS&null;n2Vo&null;&null;Vin(t)&null;.

&null;0046&null; Interval 3 (t2&null;t3): At t2, i1 reaches zero and the input rectifier bridge 140 prevents it from going negative. And iD2, which only consists of magnetizing current of T2 220 in this interval, continues to decrease until the switch is turned on at t3. At t&null;t3&null;to&null;TS, the switching cycle repeats.

&null;0047&null; C. Features

&null;0048&null; By adding another discharging path to PFC inductor, the following benefits occur:

&null;0049&null; a) The maximum intermediate bus voltage is limited. Only at boost mode when input voltage is higher than VCS&null;n2Vo&null;n1Vo, intermediate bus capacitor CS 180 is charged by input power. The higher the intermediate bus voltage, the less charging power. So the maximum intermediate bus voltage will be limited to Vin,peak&null;n1Vo&null;n2Vo. Carefully selecting transformer turn ratio n1 and n2, the maximum intermediate bus voltage can be set to a little higher than the peak value of input voltage to achieve low voltage stresses and high power factor. For universal voltage (85&null;265VAC,RMS) applications, the maximum intermediate bus voltage can be controlled to less than 400VDC, allowing single commercial 450VDC capacitor to be used in this topology. Since the maximum intermediate bus voltage is limited, DC/DC conversion cell can operate in CCM for low current stresses, without problem of high voltage at light load existing in the conventional single-stage PFC converters.

&null;0050&null; b) A portion of the load power is processed by the main switch only once. In the flyback mode, all input power is directly transferred to load by T1 120. In the boost mode, some input power is directly transferred to the load by T2 220, and some input power is stored in intermediate bus capacitor 240 and then delivered to the load 280 by the DC/DC cell. So the total power processed by active switch 200 is less than that in conventional single-stage PFC converter.

&null;0051&null; In FIG. 5, the power flows of Bi-flyback topology are illustrated. There are two directly transferred power portions. When instantaneous input voltage is low and Bi-flyback operate under flyback mode, all input power is transferred to output directly by T1 120. So P11 will be equal to input power. When the circuit enters boost mode, some input power is transferred to output by T1 120. The sum of the power provided by T1 120 and T2 220 should be equal to output power, so the power provided by T2 220 will be equal to the difference between output power and directly transferred power.

EXPERIMENTAL RESULTS

&null;0052&null; One prototype based on the topology shown in FIG. 1(c) has been built and tested to verify its operation principle. The main specifications were:

&null;0053&null; Input: approximately 85 - approximately 265V AC'RMS

&null;0054&null; Output: approximately 28VDC &null; approximately 150W

&null;0055&null; Switching frequency: approximately 200 kHz

&null;0056&null; T1 120: primary inductance L1&null;30 &null;H, turn ratio n1&null;4

&null;0057&null; T2 220: primary inductance L2&null;approximately 375 &null;H, turn ratio n2&null;approximately 3.8

&null;0058&null; Rectifier Input Bridge 140: line voltage rectifier rated at approximately 600 V. D1 160: approximately 200V/10 Amps fast recovery diode

&null;0059&null; D2 260: approximately 200V/10 Amps fast recovery diode

&null;0060&null; Power MOSFET s 200: approximately 600V/10 Amps MOSFET

&null;0061&null; Intermediate capacitor cs 240: approximately 450V/approximately 150 &null;F Aluminum capacitor

&null;0062&null; Output capacitor co 180: approximately 450V/approximately 1000 &null;F Aluminum capacitor

&null;0063&null; Load 280: approximately 150 w DC load.

&null;0064&null; FIG. 6 shows experimental waveforms of input current and voltage at 100 W output and 110V AC.RMS input.

&null;0065&null; FIG. 7 shows the variation of intermediate bus voltage versus output power at different input voltage conditions. The maximum voltage across intermediate capacitor is about approximately 390VDC for universal input voltage and al&null; load condition.

&null;0066&null; FIGS. 8 and 9 gives the measured power factor and efficiency versus input voltage at 150 W load. Measured power factor is approximately 0.974 with approximately 83.2% efficiency at approximately 110V input and approximately 150 W load.

&null;0067&null; It would be useful to now list the key features of the invention which include:

&null;0068&null; 1. Reduced cost and improved reliability due to least components;

&null;0069&null; 2. Reduced current stress. It can be shown from iL1 in FIG. 3(a), the existence of interval M3 can effectively reduce the peak current value of main switch when transferring the same average output current. Therefore, the main switch can be turned off under reduced current stress;

&null;0070&null; 3. Reduced voltage stress. Since storage capacitor is charged under the governing equation: Vcs<Vin(t)&null;n1V0&null;n2V0, therefore, using a flyback transformer to replace the traditional input inductor brings about inherent DC Bus voltage clamping capability. Properly selected n1 and n2 can guarantee the DC bus voltage well above the peak value of the input voltage, as a consequence, the commercial available 450VDC capacitor can be used, and moreover, approximately 600V components can be used in power stage for universal input applications;

&null;0071&null; 4. Direct energy transfer. In discharge mode, the input current is directly delivered to the output through the input flyback transformer T1. In charge mode, partial power is further directly transferred to the output through T1 and T2 without storing in CS first;

&null;0072&null; 5. Higher efficiency. More than half energy transferred to the load without processed twice undoubtedly can increase the overall efficiency. For the currently existed cascade two-stage or single stage approaches, basically, the power is processed serially by PFC cell and DC/DC cell, the overall efficiency is given by the product of two stage efficiencies, i.e., &null;&null;&null;1&null;2, where &null;1 and &null;2 are the efficiencies of two stages respectively. In the proposed topology, supposing &null; is the ratio at which power is transferred to the output just through PFC stage. Then, the efficiency of the proposed structure can be expressed as &null;&null;&null;&null;1&null;(1&null;&null;)&null;1&null;2>&null;1&null;2. Obviously the overall efficiency can be improved by minimizing the power process times. In addition, reduced current stress also brings about higher efficiency due to reduced turn off losses;

&null;0073&null; 6. Low turn off spikes. The snubber capacitor C1 can effectively suppress the turn off spikes of the main switch, and in each switching cycle, its stored energy can also be released to the output through the coupling winding at the moment of main switch being turn on; and,

&null;0074&null; 7. High power application. Two flyback transformers configuration has the potential to increase the power conversion rating and also release the thermal design difficulties due to distributed heat dissipation.

&null;0075&null; While the invention has been described, disclosed, illustrated and shown in various terms of certain embodiments or modifications which it has presumed in practice, the scope of the invention is not intended to be, nor should it be deemed to be, limited thereby and such other modifications or embodiments as may be suggested by the teachings herein are particularly reserved especially as they fall within the breadth and scope of the claims here appended.