CROSS-REFERENCE TO RELATED APPLICATION

This application is based upon and claims the benefit of priority from Japanese Patent Application No. 2012-201966, filed on Sep. 13, 2012, the entire contents of which are incorporated herein by reference.

TECHNICAL FIELD

The present disclosure relates to a switching regulator and a control circuit thereof.

BACKGROUND

Recent electronic apparatuses such as personal digital assistants (PDAs) and so on are equipped with devices requiring a power source voltage higher or lower than a battery voltage. A step-up, step-down or step-up/step-down type switching regulator is used to supply a proper power source voltage to such devices.

FIG. 1 is a circuit diagram showing a switching regulator 4r. The switching regulator 4r includes a control circuit 100r and an output circuit 102. The output circuit 102 includes a switching transistor M1, a synchronous rectifying transistor M2, an inductor L1 and an output capacitor C and has a topology of a step-down type switching regulator. The control circuit 100r switches the switching transistor M1 and the synchronous rectifying transistor M2 to stabilize an output voltage V_OUT to a target value.

The control circuit 100r includes a bottom detection comparator 10, a driving circuit 20r, a peak current detector 50 and a zero current detector 60. A first voltage-dividing resistor R1 and a second voltage-dividing resistor R2 divide the output voltage V_OUT to generate a feedback voltage V_FB based on the output voltage V_OUT. The bottom detection comparator 10 compares the feedback voltage V_FB with a predetermined reference voltage V_REF and generates an on signal S_ON asserted (for example, having a high level) when the feedback voltage V_FB decreases to the reference voltage V_REF.

The peak current detector 50 includes a current detector 52 and a peak current detection comparator 54 and generates an off signal S_OFF asserted when current I_M1 flowing into the switching transistor M1 reaches a predetermined peak current I_PEAK.

In an on period of the switching transistor M1, a voltage V_LX of a junction point (a switching terminal LX) of the switching transistor M1 and the synchronous rectifying transistor M2 is given by V_DD−I_M1×R_ON1. Where, R_ON1 denotes an on resistor of the switching transistor M1. The current detector 52 generates a detection voltage V_IM1 depending on a voltage drop (I_M1×R_ON1) of the switching transistor M1. The peak current detection comparator 54 compares the detection voltage V_IM1 with a threshold voltage V_PEAK corresponding to the peak current I_PEAK and asserts the off signal S_OFF (for example, having a high level) when the detection voltage V_IM1 reaches the threshold voltage V_PEAK, in other words, when the current I_M1 reaches the predetermined peak current I_PEAK.

The zero current detector 60 generates a zero current detection signal S_ZERO asserted when current I_M2 flowing into the synchronous rectifying transistor M2 decreases to a near-zero threshold value I_ZERO. In an on period of the synchronous rectifying transistor M2, a voltage V_LX of the switching terminal LX is given by V_IM2=−R_ON2×I_m2. Where, R_ON2 denotes an on resistor of the synchronous rectifying transistor M2. The zero current detector 60 includes a comparator to compare the voltage V_LX of the switching terminal LX with a predetermined threshold voltage V_ZERO.

The driving circuit 20r includes a control logic part 22r and a pre-driver 24. The control logic part 22r receives the on signal S_ON, the off signal S_OFF and the zero current detection signal S_ZERO and generates a control signal to direct turning-on/off of the switching transistor M1 and the synchronous rectifying transistor M2. The pre-driver 24 controls the switching transistor M1 and the synchronous rectifying transistor M2 based on the control signal generated by the control logic part 22r.

When the on signal S_ON is asserted, the driving circuit 20r turns on the switching transistor M1 and turns off the synchronous rectifying transistor M2. Subsequently, when the off signal S_OFF is asserted, the driving circuit 20r turns off the switching transistor M1 and turns on the synchronous rectifying transistor M2. Subsequently, when the zero current detection signal S_ZERO is asserted, the driving circuit 20r turns off both of the switching transistor Ml and the synchronous rectifying transistor M2.

FIG. 2 is an operation waveform diagram of the switching regulator 4r of FIG. 1. The bottom detection comparator 10 of FIG. 1 is configured to have a high response speed and a very small delay. At time t1, when the feedback voltage V_FB decreases to the reference voltage V_REF, the on signal S_ON is immediately asserted and the switching transistor M1 is turned on.

When the switching transistor M1 is turned on, the voltage V_LX of the switching terminal LX rises to the proximity of an input voltage V_DD. In addition, as coil current L_COIL increases, i.e., as the current L_M1 of the switching transistor M1 increases, a voltage drop of the switching transistor M1 increases and the voltage V_LX of the switching terminal LX is being lowered.

At time t2, the voltage drop of the switching transistor M1 reaches the threshold value V_PEAK. In other words, the voltage V_LX of the switching terminal LX decreases to V_DD-V_PEAK.

The peak current detector 50 has response delays, specifically, a delay in the current detector 52 and a delay in the peak current detection comparator 54. Due to the sum τ_D of these delays, the off signal S_OFF is asserted at time t3 after lapse of the delay time τ_D from time t2. At time t3, the switching transistor M1 is turned off and the synchronous rectifying transistor M2 is turned on. At time t4, when current flowing into the synchronous rectifying transistor M2 decreases to the near-zero threshold value I_ZERO, both of the switching transistor M1 and the synchronous rectifying transistor M2 are turned off.

In the switching regulator 4r of FIG. 1, for the delay time τ_D between time t2 and time t3, the coil current I_COIL, i.e., the current I_M1 of the switching transistor M1, continues to increase and reaches I_PEAK′. In other words, in order to set the actual peak current I_PEAK′ of the coil current I_COIL to a desired target value, there is a need to determine the peak current I_PEAK, i.e., a threshold voltage V_PEAK, in consideration of a length of the delay time τ_D and a slope of the coil current I_COIL.

For example, assuming that V_DD=3.7V, V_OUT=1.7V, inductance of the inductor L1 is 2.2 μH, I_PEAK′=100 mA, and τ_D=60 ns, the desired I_PEAK is approximately 45.45 mA. Assuming that R_ON1=0.2Ω, a voltage drop of the switching transistor M1 is 9.09 mV=0.2Ω×45.45 mA. That is, the peak current detector 50 needs to compare the voltage drop, which is likely to be a few millivolts (mV), with the threshold voltage V_PEAK. However, in actuality, it is difficult to compare the voltage drop in millivolts (mV) with the threshold voltage, which may result in erroneous detection.

SUMMARY

The present disclosure provides various embodiments of a switching regulator which is capable of controlling a peak of coil current with high precision.

According to one embodiment of the present disclosure, there is provided a control circuit for controlling a switching transistor and a synchronous rectifying transistor of a switching regulator. The control circuit includes: a bottom detection comparator configured to assert an on signal when a feedback voltage depending on an output voltage of the switching regulator decreases to a predetermined reference voltage; a timer circuit configured to generate an off signal asserted after lapse of on-time having a length depending on an input voltage and the output voltage of the switching regulator after the on signal is asserted; a zero current detector configured to assert a zero current detection signal when current flowing into the synchronous rectifying transistor decreases to a predetermined threshold value after the off signal is asserted; and a driving circuit configured to receive the on signal, the off signal and the zero current detection signal, and (i) turn on the switching transistor and turn off the synchronous rectifying transistor when the on signal is asserted, (ii) turn off the switching transistor and turn on the synchronous rectifying transistor when the off signal is asserted, and (iii) turn off the switching transistor and the synchronous rectifying transistor when the zero current detection signal is asserted.

With this configuration, since the off signal is generated on the basis of a time measurement by the timer circuit, rather than voltage comparison by a comparator, a peak of the coil current can be controlled with high precision.

The timer circuit may adjust the length of the on-time based on a difference between the input voltage and the output voltage. A slope of the current flowing into the switching transistor, i.e., the coil current, is proportional to a voltage across an inductor (coil) of the switching regulator, i.e., a difference between the input voltage and the output voltage. With this configuration, by changing the on-time in response to the slope of the coil current, even in a situation where at least one of the input voltage and the output voltage is varied, a peak of the coil current can remain constant.

The timer circuit may make the length of the on-time shorter as a difference between the input voltage and the output voltage becomes larger.

The timer circuit may adjust the length of the on-time to be substantially inversely proportional to a difference between the input voltage and the output voltage.

The timer circuit may include: a capacitor; a charging circuit configured to charge the capacitor with current depending on a difference between the input voltage and the output voltage; and a comparator configured to compare a voltage of the capacitor with a predetermined threshold voltage.

The charging circuit may charge the capacitor with current proportional to the difference between the input voltage and the output voltage.

The charging circuit may include a transconductance amplifier having a first input terminal receiving the input voltage and a second input terminal receiving the output voltage.

The charging circuit may include: a reference current source configured to generate reference current; a first transistor which is a P channel MOSFET (Metal Oxide Semiconductor Field Effect Transistor) having a drain electrode connected to the reference current source and a source electrode to which the input voltage is applied; a second transistor which is a P channel MOSFET having a source electrode to which the input voltage is applied, a gate electrode connected to a gate electrode of the first transistor, and a drain electrode connected to an output terminal of the charging circuit; a third transistor which is a P channel MOSFET having a source electrode to which the input voltage is applied, and a gate electrode connected to the gate electrode of the first transistor; and a fourth transistor having a source electrode connected to a drain electrode of the third transistor, a gate electrode to which the output voltage is applied, and a drain electrode connected to the output terminal of the charging circuit.

The timer circuit may include: a capacitor; a charging circuit configured to charge the capacitor with a predetermined current; a variable voltage source configured to generate a threshold voltage depending on a difference between the input voltage and the output voltage; and a comparator configured to compare a voltage of the capacitor with the threshold voltage.

The variable voltage source may generate the threshold voltage substantially inversely proportional to the difference between the input voltage and the output voltage.

The switching regulator may be of a step-down type.

The control circuit is integrated on a single semiconductor substrate. The term “integration” may include a case where all circuit elements are formed on a single semiconductor substrate, a case where some main circuit elements are integrated on a single semiconductor, and a case where some resistors, capacitors and so on are formed out of a semiconductor substrate. When circuits are integrated into a single IC, a circuit area can be reduced and characteristics of circuit elements can be uniformly maintained.

According to another embodiment of the present disclosure, there is provided a switching regulator comprising the above-described control circuit.

According to another embodiment of the present disclosure, there is provided an electronic apparatus comprising the above-described switching regulator.

Other aspects of the present disclosures may include any combinations of the above-described elements or conversion of expression of the present disclosure between methods, apparatuses and so on.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram showing a switching regulator.

FIG. 2 is an operation waveform diagram of the switching regulator of FIG. 1.

FIG. 3 is a circuit diagram showing a configuration of an electronic apparatus including a switching regulator according to an embodiment.

FIGS. 4A to 4D are circuit diagrams for illustrating an example configuration of a timing circuit.

FIG. 5 is an operation waveform diagram of the switching regulator of FIG. 3.

FIGS. 6A and 6B are schematic views showing an electronic apparatus equipped with the switching regulator.

DETAILED DESCRIPTION

Various embodiments of the present disclosure will now be described in detail with reference to the drawings. Throughout the drawings, the same or similar elements, members and processes are denoted by the same reference numerals and explanations of which will not be repeated. The disclosed embodiments are provided for the purpose of illustration, not limitation, of the present disclosure.

In the specification, the phrase “connection of a member A and a member B” is intended to include direct physical connection of the member A and the member B as well as indirect connection thereof via other member as long as the other member has no substantial effect on the electrical connection of the member A and the member B or has no damage to functions and effects shown by a combination of the member A and the member B. Similarly, the phrase “interposition of a member C between a member A and a member B” is intended to include direct connection of the member A and the member C or direct connection of the member B and the member C as well as indirect connection thereof via other member as long as the other member has no substantial effect on the electrical connection of the member A, the member B and the member C or has no damage to functions and effects shown by a combination of the member A, the member B and the member C.

FIG. 3 is a circuit diagram showing a configuration of an electronic apparatus 1 including a switching regulator 4 according to an embodiment.

The electronic apparatus 1 may be a tablet PC (Personal Computer), a laptop PC, a mobile terminal, a digital camera, a digital video camera, a portable audio player, a PDA (Personal Digital Assistant) or the like. The electronic apparatus 1 includes a load 2, a battery 3 and a switching regulator 4.

The load 2 includes various digital circuits, analog circuits and combinations thereof, including a CPU (Central Processing Unit), a DSP (Digital Signal Processor), a GPU (Graphics Processing Unit), a microcontroller, a memory, a liquid crystal driver, an audio circuit, etc.

The switching regulator 4 receives a power source voltage (also referred to as an input voltage) V_DD and steps down the power source voltage to generate a stabilized output voltage V_OUT which is then supplied to the load 2. The power source V_DD is supplied from the battery 3 or an external power supply (not shown).

The switching regulator 4 includes a control IC 100, an inductor L1, an output capacitor C1, a first voltage-dividing resistor R1 and a second voltage-dividing resistor R2, all of which may be integrated on a single semiconductor substrate.

The control IC 100 drives a switching transistor M1 and a synchronous rectifying transistor M2 based on a feedback voltage V_FB depending on an output signal (the output voltage V_OUT in this embodiment) of the switching regulator 4 and stabilizes the output voltage V_OUT to a desired level.

Although it is illustrated in this embodiment that the switching transistor M1 and the synchronous rectifying transistor M2 are incorporated in the control IC 100, these transistors may be discrete devices installed out of the control IC 100. The switching transistor M1 and the synchronous rectifying transistor M2 may be bipolar transistors.

An output circuit 102 including the switching transistor M1, the synchronous rectifying transistor M2, the inductor L1 and the output capacitor C1 has a topology of a synchronous rectification type switching regulator.

The output capacitor C1 is interposed between an output terminal P_OUT and a ground terminal. The inductor L1 is interposed between the output terminal P_OUT and a switching terminal P3 of the control IC 100. The input voltage V_DD is input to an input terminal P1 of the control IC 100 and a ground terminal P2 is grounded. The first voltage-dividing resistor R1 and the second voltage-dividing resistor R2 divide the output voltage V_OUT and provide a feedback voltage V_FB depending on the output voltage V_OUT to a feedback terminal P4 of the control IC 100.

The control IC 100 includes a bottom detection comparator 10, a driving circuit 20, a zero current detector 60 and a timer circuit 70 in addition to the switching transistor M1 and the synchronous rectifying transistor M2.

The switching transistor M1 and the synchronous rectifying transistor M2 are connected in series between the input terminal P1 and the ground terminal P1. A junction point LX of the switching transistor M1 and the synchronous rectifying transistor M2 is connected to the switching terminal P3.

The bottom detection comparator 10 compares the feedback voltage V_FB to a predetermined reference voltage V_REF and generates an on signal S_ON asserted (for example, having a high level) when the feedback voltage V_FB decreases to the reference voltage V_REF. The bottom detection comparator 10 is configured to have a high speed response.

The output voltage V_OUT of the switching regulator 4 is input to an output voltage detection terminal P5. The timer circuit 70 measures the on-time T_ON after the on signal S_ON is asserted, that is, after the switching transistor M1 is turned on, and generates an off signal S_OFF asserted after a lapse of the on-time T_ON. The timer circuit 70 sets a length of the on-time T_ON based on the input voltage V_DD and the output voltage V_OUT of the switching regulator 4.

The zero current detector 60 asserts a zero current detection signal S_ZERO when current IM2 (coil current I_COIL) flowing into the synchronous rectifying transistor M2 decreases to a near-zero threshold value I_ZERO after the synchronous rectifying transistor M2 is turned on with assertion of the off signal S_OFF. In an on period of the synchronous rectifying transistor M2, a voltage V_LX of the switching terminal LX is given by V_IM2=−R_ON2×I_M2. Where, R_ON2 denotes an on resistor of the synchronous rectifying transistor M2. The zero current detector 60 may include a comparator to compare the voltage V_LX of the switching terminal LX with a predetermined threshold voltage V_ZERO. In addition, the zero current detector 60 may be configured in different ways without being limited to that shown in FIG. 3. For example, a resistor for current detection may be connected in series to the synchronous rectifying transistor M2 and a voltage drop of the resistor may be compared to the threshold voltage V_ZERO.

The driving circuit 20 includes a control logic part 22 and a pre-driver 24. The control logic part 22 receives the on signal S_ON, the off signal S_OFF and the zero current detection signal S_ZERO and generates control signals S1 and S2 to control turning-on/off of the switching transistor M1 and the synchronous rectifying transistor M2. The pre-driver 24 switches the switching transistor M1 and the synchronous rectifying transistor M2 based on the control signals S1 and S2.

The driving circuit 20 (i) turns on the switching transistor M1 and turns off the synchronous rectifying transistor M2 when the on signal S_ON is asserted (a first state), (ii) turns off the switching transistor M1 and turns on the synchronous rectifying transistor M2 when the off signal S_OFF is asserted (a second state), and (iii) turns off the switching transistor M1 and the synchronous rectifying transistor M2 when the zero current detection signal S_ZERO is asserted (a third state). The driving circuit 20 repeats the first to third states and stabilizes the output voltage V_OUT to a target value depending on the reference voltage V_REF.

In the electronic apparatus 1, one or both of the input voltage V_DD and the output voltage V_OUT may be varied. For example, when the input voltage V_DD is supplied from the battery 3, the input voltage V_DD decreases as the battery 3 is discharged, whereas the input voltage V_DD increases as the battery 3 is charged. In the case of using a lithium ion battery as the battery 3, for example, the input voltage V_DD may be varied within a range of 3.2V to 4.2V.

It is also assumed that the output voltage V_OUT supplied to the load 2 may be dynamically varied.

A slope of the coil current I_COIL is proportional to a voltage across the inductor L1. During an on-period of the switching transistor M1 and an off-period of the synchronous rectifying transistor M2, the input voltage V_DD is applied to one end of the inductor L1 and the output voltage V_OUT is applied to the other end. That is, the voltage across the inductor L1 becomes V_DD-V_OUT.

In a first state, the coil current I_COIL increases with a slope ((V_DD-V_OUT)/L). The coil current I_COIL has a peak after a lapse of the on-time T_ON and the peak current I_PEAK is given by the following equation 1.

I_PEAK=(V_DD−V_OUT)/L×T_ON   [Equation 1]

When at least one of the input voltage V_DD and the output voltage V_OUT is varied, the peak current I_PEAK can remain constant by adjusting the on-time T_ON to cancel such variation. That is, the on-time T_ON is adjusted to satisfy the following equation 2.

T_ON=I_PEAK/(V_DD−V_OUT)×L   [Equation 2]

From a different standpoint, the timer circuit 70 of FIG. 3 adjusts the length of the on-time T_ON based on a difference (V_DD-V_OUT) between the input voltage V_DD and the output voltage V_OUT. The timer circuit 70 may make the length of the on-time T_ON shorter as the difference (V_DD-V_OUT) between the input voltage V_DD and the output voltage V_OUT becomes larger. In other words, the timer circuit 70 may adjust the length of the on-time T_ON to be substantially inversely proportional to the difference (V_DD-V_OUT) between the input voltage V_DD and the output voltage V_OUT.

FIGS. 4A to 4D are circuit diagrams for illustrating example configurations of the timer circuit 70. A timer circuit 70a of FIG. 4A includes a capacitor C11, a switch SW11, a current source 72 and a comparator 74. The capacitor C11 has one end having a fixed potential and the other end to which the current source 72 is connected. The current source 72 charges the capacitor C11 with charging current I_CHG depending on the input voltage V_DD and the output voltage V_OUT. The switch SW11 is provided to initialize a voltage V_C11 of the capacitor C11 to zero and is switched off as the on-time T_ON is started. The comparator 74 compares the voltage V_C11 of the capacitor C11 with a predetermined threshold voltage V_TH and generates an off signal S_OFF asserted (for example, having a high level) when the voltage V_C11 reaches the threshold voltage V_TH.

The on-time T_ON measured from when the switch SW11 is switched off until when the off signal S_OFF is asserted is given by the following equation 3.

T_ON=C11×V_TH/I_CHG   [Equation 3]

The current source 72 may generate the charging current I_CHG substantially proportional to the difference voltage (V_DD-V_OUT). Using a proportional coefficient g_m, the following equation 4 is established.

I_CHG=g_m×(V_DD−V_OUT)   [Equation 4]

The following equation 5 can be obtained by putting Equation 4 into Equation 3.

T_ON=C11×V_TH/{g_m×(V_DD−V_OUT)}  [Equation 5]

In comparison between Equation 2 and Equation 5, the peak of the coil current I_COIL can be set to a desired value I_PEAK by adjusting the capacitance of the capacitor C11, the threshold value V_TH and the proportional coefficient g_m, as can be seen from the following equation 6.

I_PEAK/L=C11×V_TH/g_m   [Equation 6]

FIG. 4B is a circuit diagram showing an example configuration of the current source 72 of FIG. 4A. The current source 72 includes a transconductance (gm) amplifier and outputs the current I_CHG depending on the voltage difference (V_DD-V_OUT), more specifically, outputs the current I_CHG substantially proportional to the voltage difference (V_DD-V_OUT). A proportional coefficient corresponds to mutual conductance g_m of the transconductance amplifier.

FIG. 4C is a circuit diagram showing another example configuration of the current source 72 of FIG. 4A. The current source 72 includes first to fourth transistors M11 to M14, which are P channel MOSFETs, and a reference current source 76. The reference current source 76 generates a predetermined reference current I_REF. The first to third transistors M11 to M13 form a current mirror circuit and have their respective source electrodes to which the input voltage V_DD is applied. The fourth transistor M14 is interposed between the third transistor M13 and an output terminal 78 of the current source 72 and has its gate electrode to which the output voltage V_OUT is applied.

Current I_M14 flowing into the fourth transistor M14 becomes larger as a difference between V_DD and V_OUT becomes larger. The current I_M14 is added to current I_M12 flowing into the second transistor M₁₂ to generate the charging current I_CHG. Although the charging current I_CHG is not completely proportional to the voltage difference between V_DD and V_OUT the charging current I_CHG can be used for the timer circuit 70a since this current depends on the voltage difference between VDD and V_OUT.

FIG. 4D shows another configuration of the timer circuit. A timer circuit 70b has the same basic configuration as the timer circuit 70a of FIG. 4A. However, a current source 72b generates a predetermined charging current I_CHG in a different way as described below. A variable voltage source 79 generates a threshold voltage V_TH based on the difference between the input voltage V_DD and the output voltage V_OUT.

The on-time T_ON measured form when the switch SW11 is switched off until when the off signal S_OFF is asserted is given by the above equation 3.

The variable voltage source 79 may generate a threshold voltage V_TH decreasing with increase in the difference voltage (V_DD-V_OUT). In other words, the threshold voltage V_TH may be substantially inversely proportional to the difference voltage (V_DD-V_OUT). Using an inverse-proportional coefficient, k_m, the following equation 7 is established.

V_TH=k_m/(V_DD−V_OUT)  [Equation 7]

The following equation 8 can be obtained by putting Equation 7 into Equation 3.

T_ON=C11×k_m/(V_DD−V_OUT)/I_CHG   [Equation 8]

In comparison between Equation 2 and Equation 8, the peak of the coil current I_COIL can be set to a desired value I_PEAK by adjusting the capacitance of the capacitor C11, the charging current I_CHG and the coefficient k_m, as can be seen from the following equation 9.

I_PEAK/L=C₁₁×k_m/I_CHG   [Equation 9]

The configuration of the variable voltage source 79 is not particularly limited but may employ circuits known or to be developed in future.

Subsequently, operation of the switching regulator 4 will be described. FIG. 5 is an operation waveform diagram of the switching regulator 4 of FIG. 3. At time t1, when the feedback voltage V_FB decreases to the reference voltage V_REF, the on signal S_ON is asserted. Due to this assertion, the switching regulator 4 transitions the first state, the switching transistor M1 is turned on and the synchronous rectifying transistor M2 is turned off.

When the switching regulator 4 transitions to the first state, the on-time T_ON is measured by the timer circuit 70 and the off signal S_OFF is asserted at time t3 after lapse of the on-time T_ON. Due to this assertion, the switching regulator 4 transitions to the second state Φ2, the switching transistor M1 is turned off and the synchronous rectifying transistor M2 is turned on.

At time t4, the zero current detection signal S_ZERO is asserted when coil current I_COIL (I_M2) flowing into the synchronous rectifying transistor M2 decreases to a near-zero threshold value I_ZERO. Due to this assertion, the switching regulator 4 transitions the second state Φ3, both of the switching transistor M1 and the synchronous rectifying transistor M2 are turned off.

The switching regulator 4 has advantages over the switching regulator 4r of FIG. 1.

In the switching regulator 4r of FIG. 1, the actual peak I_PEAK of the coil current I_COIL is affected by the delay τ_D of the peak current detector 50 generating the off signal S_OFF. That is, the input voltage V_DD and the output voltage V_OUT are varied and a slope of the coil current I_COIL for the delay time τ_D is changed depending on the input voltage V_DD and the output voltage V_OUT. Accordingly, an amount of increase of the coil current I_COIL in the delay time τ_D and further the actual peak current I_PEAK are also varied depending on the input voltage V_DD and the output voltage V_OUT. When the peak current I_PEAK is varied, a ripple width of the output voltage V_OUT is also varied to reduce stabilization of the output voltage V_OUT.

In contrast, according to the switching regulator 4 of FIG. 3, since the timer circuit 70 generates the off signal S_OFF by measuring the on-time T_ON, it is possible to exclude an effect by a delay as in the peak current detector 50. In addition, although the peak current is also changed by an offset of the peak current detection comparator 54 in the peak current detector 50, such an affect by the offset can be excluded in the switching regulator 4 of FIG. 3.

In addition, by adjusting the on-time T_ON based on the input voltage V_DD and the output voltage V_OUT, the peak current I_PEAK, i.e., a current ripple, can remain constant and a ripple amount of the output voltage V_OUT can also remain constant.

Heretofore, the present disclosure has been described by way of specific embodiments. The disclosed embodiments are merely examples and it is to be understood by those skilled in the art that combinations of elements and processes of the embodiments can be modified in various ways and such modification falls within the scope of the present disclosure. The following description is given to such modification.

(First Modification)

There is no need for the on-time T_ON to be completely inversely proportional to the difference between the input voltage V_DD and the output voltage V_OUT. Even in a deviation from such an inverse proportion relationship, by making the on-time T_ON shorter as the difference between the input voltage V_DD and the output voltage V_OUT becomes larger, variation of the peak current I_PEAK can be more effectively suppressed in comparison to fixed on-time T_ON.

(Second Modification)

Although it has been illustrated in the above embodiments that the on-time T_ON is changed depending on the input voltage V_DD and the output voltage V_OUT, the on-time T_ON may be fixed if variation of the difference between the input voltage V_DD and the output voltage V_OUT is small. Even in this case, the peak current can be correctly controlled since it is not affected by the delay time of the peak current detector 50, as opposed to the switching regulator 4r of FIG. 1.

(Third Modification)

Although a step-down type switching regulator has been illustrated in the above embodiments, the present disclosure is not limited thereto but may be applied to a step-up type or step-up/step-down type switching regulator. In such a step-up type or step-up/step-down type switching regulator, a topology of the output circuit 102 need to be changed.

Finally, an example of the electronic apparatus 1 will be described. FIGS. 6A and 6B are schematic views showing an electronic apparatus equipped with the switching regulator 4. An electronic apparatus 500 of FIG. 6A is a tablet PC, a portable game machine or a portable audio player and has a housing 502 in which the battery 3, the switching regulator 4 (the control IC 100) and the load 2 are incorporated. The load 2 is, for example, a CPU.

An electronic apparatus 600 of FIG. 6B is a digital camera and has a housing 602 in which the battery 3, the switching regulator 4, an imaging device 2a, an image processor 2b and so on are incorporated. The switching regulator 4 supplies a power source voltage to the imaging device 2a and the image processor 2b.

According to the present disclosure in some embodiments, it is possible to provide a switching regulator which is capable of controlling a peak of coil current with high precision.

While certain embodiments have been described, these embodiments have been presented by way of example only, and are not intended to limit the scope of the disclosures. Indeed, the novel methods and apparatuses described herein may be embodied in a variety of other forms; furthermore, various omissions, substitutions and changes in the form of the embodiments described herein may be made without departing from the spirit of the disclosures. The accompanying claims and their equivalents are intended to cover such forms or modifications as would fall within the scope and spirit of the disclosures.