BACKGROUND OF THE INVENTION

The present invention relates to a method for suppressing resonance phenomena in the a-c network on the inverter side of a high-voltage d-c transmission system (HVDCTS). The invention further relates to an apparatus for carrying out the method.

With reference to FIG. 1, an HVDCTS is shown which is arranged between two a-c networks NA, NB and in which a first converter 1A of the HVDCTS which is connected in a first station (STA) to the first a-c network NA, is clocked by the network and is operated as a rectifier, impresses a predetermined d-c current i_dA from the first a-c network and in a second station (STB), a second converter 1B which feeds into the second a-c network NB, is clocked by the network and is operated as an inverter, determines the level U_dB of the transmitted d-c voltage.

In the preferred embodiment shown of a short coupling, the two stations are physically close together and the HVDCTS comprises only the d-c lines 2 and 3, a converter choke LA and LB being arranged at the d-c terminal of each converter. Each station contains a regulator 4A, 4B, the output signal of which determines, as the control voltage Δx_A, Δx_B, respectively, the control angle for the control unit STA and STB of the respective converter. The control units can be supplemented in the case of a digital control unit by supplementary devices, not shown, for instance, for linearizing the characteristic and/or for converting the given analog control voltage into a digital signal in the case of a digital control unit.

The control voltage Δx_A determines the d-c voltage U_dA at the d-c terminal of the converter 1A and therefore, if the d-c current i_dA is set into the regulator 4A (current regulator), the control deviation of the d-c current fed-in by the converter 1A, the transmitted HVDCTS d-c current i_d which is equal to the d-c current i_dB which is taken from the converter 1B of the HVDCTS and is fed into the network NB, since in the case of a short coupling, power losses occurring in thyristors and chokes can be ignored (i_d =i_dA =i_dB). To the regulator 4B of station B (for instance, a voltage regulator or quenching angle regulator) can be fed, in the interest of holding the voltage of the network NB constant, the control deviation of the voltage amplitude of this network or, in the interest of an efficiency as high as possible for the power transmission and low reactive power, the control deviation of the quenching angle of the converter 1B can be pre-set by a predetermined designed quenching angle value which is a maximum as far as possible (i.e., is close to the inverter out-of-step limit). Its control voltage Δx_B determines the d-c voltage U_db associated with the impressed HVDCTS current i_d at the d-c terminals of the converter 1B and thereby, the voltage level U_d of the transmitted d-c voltage which is taken off between the two converter chokes LA and LB and, in the case of a long distance HVDCTS line, preferably at the end of the choke LA facing away from the converter 1A.

The two stations are coupled to each other via the HVDCTS in such a manner that any change of Δx_B and U_dB in station B causes, in accordance with the voltage drop at the HVDCTS, a change of U_dA and thereby, a current change in station A which must be equalized by the current regulator 4A. On the other hand, any change of Δx_A or U_dA in the station A brings about a change of i_d and U_dB in station B, to which the regulator 4B responds in station B. In order to keep interfering effects of this coupling as low as possible, it has already been proposed to impress on the output signal Δx_A of the current regulator 4A a pilot voltage U_dAv, which is taken off directly at the measuring stage for U_d or is formed by means of a computing circuit as the model variable. Likewise, a pilot voltage U_dBv can be impressed on the control voltage Δx_B, which is formed in a different manner, for instance, by calculating the inductive d-c voltage drop of the converter 1B.

For the network NB, the HVDCTS operates in practice so that the voltage oscillations are reflected at the converter, wherein a small ripple of U_d and i_d is produced which is equalized by the current regulator 4A if this current regulator works fast enough. In FIG. 1, it is indicated that the network NB can contain a disturbing voltage oscillation which in this manner acts like an oscillation coupled into the HVDCTS circuit via an interference quantity U_B for the control angle Δx_B with a generally changed amplitude, phase and frequency.

If in the network, this interfering oscillation is located in the vicinity of the resonance frequency of the network and if the current regulator 4A is not able to equalize the resonance frequency generated in the d-c circuit, this interfering oscillation can thereby be fanned-up in such a manner that considerable ripple of U_d and i_d in the HVDCTS circuit comes about, in which the d-c voltage corresponding to the nominal voltage provided is superimposed by a "resonance oscillation" which can exceed, for instance, 30% of the nominal voltage and can ultimately lead to the situation that the entire system must be shut down. This is the case, for instance, in a specific case, in which the resonance frequency of the network NB is in the vicinity of the second harmonic of the network NB, it having been found that the interfering resonance frequency of the d-c circuit is approximately equal to the fundamental (60 Hz) of the 3-phase network NB.

SUMMARY OF THE INVENTION

It is therefore an object of the present invention to suppress such a resonance phenomenon in an HVDCTS between the a-c voltage on the inverter side and the transmitted d-c voltage or the transmitted d-c current.

The basic idea of the invention is to determine the resonance phenomenon (in the example, therefore, the 60Hz oscillation) in the d-c circuit and to attenuate it by intervention into the control of the rectifier station. Since the control 4A of this station is as a rule, too sluggish for this purpose, a control variable corresponding to the resonance phenomenon is applied as a pilot voltage ΔU_A of (possibly already servo-controlled by the pilot voltage U_dAv) control voltage Δx_A at the output of the regulator 4A. While thereby the resonance oscillation in the d-c circuit is ultimately also coupled into the network NA, the resonance frequency of the network NA is almost always different from the resonance frequency of the network NB, and in addition, the phase and frequency of the network NA always changes somewhat relative to the network NB, further resonance phenomena in the network NA do not come about, while the resonance of the network NB can be damped effectively.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be explained in greater detail, in the following detailed description with reference to the drawings, in which:

FIG. 1 is a block diagram of the present invention; and

FIG. 2 is a graph showing signals present in the diagram of FIG. 1.

DETAILED DESCRIPTION

If the operating point of station B, particularly the interfering frequency and the voltage U_dB (or the voltage U_d or the current i_d caused thereby) are known well enough and if the actual value of the HVDCTS voltage U_d is determined without delay and the change of the control angle of the first converter can be adjusted without delay, it would be sufficient to filter out ("isolate") the resonance oscillation component (in the example, therefore, the 60Hz oscillation) from the HVDCTS voltage U_d without phase shift. If then the isolated oscillation component is added to the control voltage with amplification tuned to the operating point, it can be achieved thereby that the output voltage U_dA is changed proportionally to and in phase with the HVDCTS volta U_d and a constant i_d =i_dA is thereby impressed on the choke LA. The HVDCTS then reflects for all practical purposes no longer the interference oscillation contained in the network NB but it becomes permeable therefor and the cause of the resonance oscillation is thereby largely eliminated and the resonance is damped.

In principle, the same can be achieved if the resonance oscillation component of the HVDCTS current i_d is filtered out and is added as a pilot variable to Δx_A. However, the current i_d follows the change generated by a periodic change of the control angle controlling the converter 1A with a respective phase shift of 90° so that a phase rotation of 90° can be produced between the pilot quantity derived from i_d and the change of the control angle due to the pilot quantity.

The idealizing conditions (particularly determination and adjustment of the voltage U_d without delay and filtering out the resonance oscillation component without phase rotation) do not exist, however, in reality. Thus the measuring transformers which are required for determining and stepping down the high voltage U_d have a considerable time constant which is frequently increased still further by additional smoothing. Also the control unit of the converter 1A exhibits a switching delay and additional smoothing. The control circuit which leads from the measuring transformer for determining the oscillation component to the actual value of the oscillation component via the addition of the pilot variable derived from the oscillation component, therefore has a time constant ("time constant of the pilot") which must be taken into consideration in order to produce the desired phase equality of U_dA and U_d.

On the other hand, filtering out the oscillation component which is superimposed on the d-c voltage or the d-c current in the HVDCTS d-c circuit is as a rule possible only with phase rotation. This, however, can be utilized for equalizing through appropriate design of the required filter circuit, just that time constant of the pilot, i.e., the phase angle belonging to this time constant and to the frequency of the resonance oscillation. The filter circuit for determining the measured value is therefore tuned to this phase angle φ. Thereby, however, heavy damping is possible only for a given operating point. If therefore only one pilot quantity is formed, the design of the filter circuit (phase rotation and gain) must follow the change of the operating point in the event of a change of the operating point in order to obtain optimum damping also in the case of a change of the amplitude of U_d or i_d or of the frequency of the resonance oscillation.

If, in addition, the pilot quantity is derived from the d-c current i_d, it must be added that this pilot quantity damps just that oscillation of i_d which is required for forming the pilot quantity itself. Thus, the better the damping is to be, so much more difficult becomes the determination of the damping pilot quantity.

These difficulties are avoided if both measures are combined with each other. It is then sufficient to use for the formation of the control input derived from the measured voltage value a filter circuit which is tuned to a central operating point. Any deviation of the actual operating point from this mean operating point then means a detuning of the resonance conditions corresponding to overcompensation or undercompensation of the resonance oscillation, from which a resonance oscillation in the measured current value follows. From this current resonance can then be formed a further pilot quantity, with gain, by a phase rotation which amounts to about 90°. By the addition of the total pilot quantity resulting as the sum of both pilot quantities, an optimum damping for operating points variable within a wide operating range can be achieved with constant lay-out of the circuit.

In this connection it has been found that the tuning of the filter circuit for the measured current value is not very critical. In particular, it is frequently not necessary to take also into consideration the phase angle due to the mentioned time constant of the pilot for the phase rotation of the current resonance oscillation component beyond the 90° phase shift.

In the embodiment of FIG. 1, the filter circuit 10 is realized by a bandpass filter 101 tuned to the mean resonance frequency of the d-c circuit, i.e., 60 Hz, with a following amplification stage 102. This bandpass filter accordingly isolates this frequency of the d-c circuit from the d-c component and the harmonics caused by the converter operation, and brings about at the same time the desired phase shift by the angle φ.

Any deviation from the operating point on which the filter circuit 10 is based makes it necessary that the current i_d has ripple. Depending on how, for instance, the d-c voltage U_d or the frequency of NB is changed, overcompensation or undercompensation of the resonance of the network can result by the output voltage U'_d of the filter circuit 10.

It is therefore provided advantageously to form also a quantity i'_d determining the d-c current i_d and to impress the control voltage Δx_A in such a manner that thereby the over-or undercompensated oscillation of the d-c voltage U_d is coupled with positive or negative feedback to an additional oscillation corresponding to the current resonance.

For this purpose, also the resonance oscillation contained in the measured value i_d of the HVDCTS is isolated with a phase rotation of about 90°, is amplified suitably and is impressed on the control voltage Δx_A together with U'_d. Filtering out the current resonance frequency and its phase rotation is preferably realized in the filter circuit 11 of the arrangement by forming the derivative d/dt (i_d) of the d-c current, where the differentiator 111 can be followed by a smoothing member 112 with little smoothing. Furthermore, an amplifier 113 is connected thereto. The two filter circuits 10 and 11 are tuned to each other and to the operating range of the HVDCTS in such a manner that optimum smoothing is achieved practically for all occurring operating points.

According to FIG. 1, it is further provided advantageously to execute the damping addition of the pilot quantity U_A only if a disturbing resonance occurs in the network NB and thereby, a noticeable resonance in the d-c circuit. Therefore, a filter tuned to the interfering resonance frequency (in the example, 60 Hz) of the d-c circuit for the measurement value of U_d (or preferably of i_d) is provided which isolates the resonance frequency, forms its amplitude via a rectifier 122 and feeds the latter to a threshold member 123. If the resonance amplitude exceeds a predetermined critical threshold value, a time delay member 124 is triggered which drives a transistor switch 125 into conduction. The switch 125 then adds during this intervention time the damping pilot quantity ΔU_A to the control voltage, while the resonance phenomenon in the network NB is damped and also disappears quickly completely, depending on the network configuration.

In FIG. 2, three states of the device according to FlG. 1 are shown. First, state I will be considered, in which the two networks NA and NB contain only the respective fundamental and their voltage amplitudes are therefore proportional to the control voltage Δx_A and Δx_B and are therefore constant. In this undisturbed case, also the d-c voltages U_d, U_A and U_dB as well as the d-c current i_d and the transmitted active power P_d are constant.

If now the resonance frequency of the network NB is equal to the second harmonic of this a-c voltage network, a corresponding oscillation is excited and is fanned-up to a considerable amplitude. This resonance oscillation generates in the measured value U_d of the d-c voltage a superimposed resonance frequency with the frequency of the fundamental, which can be simulated, for instance, by the provision that the corresponding resonance oscillation ΔU_B is superimposed on the control angle of the control unit 1B. With the occurrence of the resonance in the network NB, the HVDCTS changes therefore from the undisturbed state I to the resonance state II.

The current regulator 4A cannot compensate this oscillation and remains practically at its constant value. Consequently, also the d-c output voltage U_dA is constant, while the d-c voltage U_dB and U_d exhibit the oscillation generated by ΔU_B. Corresponding to the voltage difference U_dA -U_dB, a resonance frequency is also produced in the d-c current i_d, so that the transmitted active power P_d contains considerable pulsations. The HVDCTS therefore becomes permeable to a certain extent to the resonance phenomena of the network NB which are ultimately coupled into the network NA and are damped only there.

If now the pilot control variable ΔU_A is formed from U_d and i_d, the d-c voltage U_dA is now proportional to Δx_A +ΔU_A, where U_A is now determined so that the driving voltage U_dB -U_dA in the HVDCTS becomes constant and thereby generates a constant d-c current i_d.

By keeping the HVDCTS current constant in this manner, the permeability of the HVDCTS and therefore the damping of the resonance in NB is greatly increased and, after the switch 125 is closed, changes from state II quickly back to state I.

To better illustrate the action of the pilot quantity ΔU_A better, an operating state is shown by III in FIG. 2, in which the state with the excited resonance frequency is maintained artificially by adding a further control variable ΔU_B to the output of the regulator 4B in the network NB. With this simulation it can be seen clearly that in state III, the d-c voltage U_dB, U_d and U_dA are proportional and of the same phase so that the desired constancy of the d-c current is assured. However, this is accompanied by a pulsation of the active power P_d, where now the added pilot control quantity ΔU_A couples a certain amount of harmonics into the network NA. Since, however, state III, in which the resonance in the network NB has a considerable amplitude and the pilot control quantity ΔU_A is added, occurs only immediately after the switch 125 is closed and thus state III is assumed to be only temporary during the change from state II to state I, the pulsation of the active power and the interference frequency coupled into the network NA are practically without importance.

It becomes possible thereby to suppress a resonance phenomenon in the network NB by appropriate intervention into the control of the rectifier by practically eliminating the cause, namely, the reflection of the interference oscillation.

In the foregoing specification, the invention has been described with reference to a specific exemplary embodiment thereof. It will, however, be evident that various modifications and changes may be made thereunto without departing from the broader spirit and scope of the invention as set forth in the appended claims. The specification and drawings are, accordingly, to be regarded in an illustrative rather than in a restrictive sense.